The invention relates to a blocking oscillator switched mode power supply for supplying electrical equipment, wherein the primary winding of a transformer in series with the current-carrying section of an electronic switch is connected to the DC voltage obtained by rectification of the power AC voltage supplied via two supply terminals, and a secondary winding of the transformer is provided to supply current to the electrical equipment, wherein furthermore the control electrode of the three terminal electronic switch is controlled by the output of a control circuit which in turn is actuated by the rectified power AC voltage as actual value and by a set point adjuster, wherein further a starting circuit is provided for the further control of the control electrode of the electronic switch, and wherein lastly the control circuit is constructed so that its current supply is given by means of a secondary winding of the transformer which in turn contains on the one hand a circuit part serving for control voltage generation, with following variable gain amplifier, and on the other hand a circuit part for pulse processing, the output of the variable gain amplifier as well as the output of the pulse processing system being each connected to an input of a pulse duration modulator actuating the control electrode of the electronic switch and forming the output of the control circuit, while a third input of the pulse duration modulator is actuated by a current-voltage converter.
Such a blocking oscillator switched mode power supply is described for example in DE-OS 30 32 034. As additional art may be cited "Funkschau (1975), No. 5, p. 40-43", or the book by Wuestehube et al. titled "Schaltnetzteile" (Switched Mode Power Supplies) (published 1979 in expert-Verlag, VDE-Verlag (cf. in particular p. 87 ff) or Siemens "Schaltnetzteile mit der IS TDA 4600" p. 7 ff.
As is known, such a switched mode power supply supplies electronic apparatus, e.g. a television receiver, with stabilized and regulated operating voltages. The core of such a switched mode power supply, therefore, is a control circuit, the positioning element of which is constituted by the initially mentioned three terminal electronic switch, realized in particular by a bipolar power transistor. Further there is provided a high working frequency and a transformer aligned to a high operating frequency, since it is generally desired to isolate of the electrical equipment to be supplied from the supply network is desired. Such switched mode power supplies may be laid out either according to the synchronized mode or according to the self-heterodyning mode. The latter applies to a switched mode power supply as described in DE-OS 30 32 034 and which the present invention deals with also.
The basic circuit diagram belonging to such a switched mode power supply is illustrated in FIG. 1, which will be discussed initially.
An npn power transistor T serves as a control element and is driven by the control circuit RS and is connected by its emitter-collector path in series with the primary winding W.sub.p of a transformer Tr. The control element could alternatively be another three terminal device, e.g. a thyristor or a power MOS field effect transistor. With reference to FIG. 1 in DE-OS 30 32 034, it can be noted that the DC voltage operating this series connection is obtained by rectification of the AC voltage supplied by the AC network by means of a rectifier circuit, e.g. a Graetz circuit. When an npn transistor is used as the control element, the emitter of this transistor is connected to ground, the collector to the primary winding W.sub.p of this transformer Tr, and the other end of this primary winding to the supply potential +U.sub.p supplied by the rectifier circuit mentioned (but not shown in the drawing). The emitter-collector path of transistor T is bridged by a capacitor Cs, while the capacitance C.sub.w at the primary winding W.sub.p is of a parasitic nature. The power transistor T is controlled at its base by the output part of the initially mentioned control circuit RS, i.e. by the pulse duration modulator PDM provided therein. When using a three terminal device other than a transistor, the control electrode is connected to correspond with the base electrode, the current input lead electrode is connected to correspond with the emitter connection, and the current output electrode is connected to correspond to the collector of the npn transistor T shown in FIG. 1.
An auxiliary winding W.sub.H of the transformer Tr serves as a sensor for the control circuit RS and therefore is connected at one end to ground and at the other end to the input of the control circuit RS. An additional secondary winding W.sub.s forms the actual secondary side of the blocking oscillator transformer Tr, which is provided for the actuation of the rectifier system GL, and the latter for the actuation of a load R.sub.L. The DC voltage supplied by the rectifier system GL is hereafter designated by U.sub.s.
The control circuit RS contains the output circuit part PDM which controls the transformer T and is designed as a pulse duration modulator, and also contains two input parts controlled by the auxiliary winding W.sub.H, one input part RSE serving for control voltage generation and supplying via a control amplifier RV a control signal U.sub.A for the output part PDM. The other input part IAB serves for pulse processing and supplies a signal U.sub.N to the output part PDM of the control circuit RS. Lastly there is provided a current-voltage converter SSW, which forms the actual value control of the control circuit and supplies a voltage U.sub.Ip proportional to the primary current I.sub.p to the pulse duration modulator PDM. The last-named parts of the control circuit RS are indicated in the DE-OS 30 32 034 also. They belong to the control circuit illustrated there in FIG. 3. The control voltage generation is effected by the resistors R5 and R4 appearing there in FIGS. 1 and 2. The pulse processor IAB consists, as shown in FIG. 3 of this reference, of a zero crossing identification and the control logic actuated by it. The pulse duration modulator PDM lastly is constituted by the trigger circuit indicated in DE-OS 30 32 034 with the part of the control logic actuated by it.
In FIG. 2 is shown the time diagram belonging to a circuit according to FIG. 1, that is, the time response of the signals occurring in the control circuit RS, namely U.sub.H (=signal supplied by the transformer winding for controlling the control circuit), U.sub.N (=signal supplied by the pulse processor IAB), I.sub.p (=current supplied by the transformer winding W.sub.p in series with the switching transistor T) and U.sub.Ip (=the actual-value signal supplied by the current-voltage converter SSW).
As is evident, the voltage U.sub.H supplied by the transformer winding W.sub.H supplies with the zero crossing (U.sub.H =0 V) the information that the energy stored in transformer Tr has drained and a new charging cycle can begin, i.e. the switch constituted by transistor T can be closed. This information is communicated to the pulse duration modulator PDM via the pulse processing stage IAB. (There applies in this case: U.sub.N &lt;0 V.fwdarw.pulse start, U.sub.N &gt;0 V.fwdarw.pulse start not possible.)
Further, with the aid of the control voltage generator RSE, there is obtained from the signal voltage U.sub.H supplied by the winding W.sub.H of transformer Tr a control voltage U.sub.R proportional to the secondary voltage U.sub.s. In the variable gain amplifier RV, the control voltage is compared with a reference. The difference between the control voltage U.sub.R and the reference is amplified by the variable gain amplifier and communicated by the signal voltage U.sub.A, which is supplied by the output of the variable gain amplifier RV, to the pulse duration modulator PDM, which compares it with the signal U.sub.Ip of the current-voltage converter SSW and opens the switch constituted by transistor T as soon as U.sub.Ip .gtoreq.U.sub.A. In this manner the peak value I.sub.pmax of I.sub.p is corrected until the difference between U.sub.R and the reference voltage disappears. This means that U.sub.R and hence U.sub.s remain constant.
As has been mentioned above, FIG. 2 illustrates important signals of the circuit according to FIG. 1. Concerning the theory, reference can be made to the cited book by Wuestehube.
In practice, a switched mode power supply must be short circuit proof with respect to U.sub.s. This means that as a certain value I.sub.smax for the secondary direct current is exceeded, the secondary voltage U.sub.s collapses. For the operation of the switched mode power supply, therefore, a power limitation is required.
In most cases the output voltage U.sub.A of the variable gain amplifier RV is limited for this purpose or even controlled back (formation of an overload diagram).
Taking first the simple case of power limitation, the maximum secondary power P.sub.smax obeys the equation EQU P.sub.smax =U.sub.s .multidot.I.sub.smax =(.mu..multidot.W.sub.pmax)/T.sub.max ( 1)
where I.sub.smax =maximum secondary current
.mu.=efficiency PA1 W.sub.pmax =the primary energy absorbed and PA1 T.sub.max =period duration of the switched mode power supply oscillation when the secondary current Is has the value I.sub.smax. PA1 1. on the value of U.sub.Amax or PA1 2. on the value of .tau..sub.ssw or PA1 3. on both U.sub.Amax and .tau..sub.ssw
Further consideration shows that I.sub.smax depends on the primary voltage U.sub.p. In fact, if the switched mode power supply is rated for a certain range of the primary voltage U.sub.p, the transformer Tr of the oscillator must be dimensioned for power ratings which however, in unfavorable cases, cannot be fully utilized at all. Thus for instance in the voltage range 90 V.ltoreq.U.sub.p .ltoreq.270 V the switched mode supply must be provided with a transformer Tr which must be overdimensioned approximately 100% in order to be able to deliver a certain power in the entire voltage range.
Using equation (1) together with the further equations EQU W.sub.pmax =(L.sub.p .multidot.I.sub.pmax.sup.2)/2 (2)
(I.sub.pmax =peak value of current I.sub.p at a pulse width t.sub.p, which equals the maximum primary pulse width t.sub.pmax for the pulse applied to the control electrode of the setting member T at a voltage U.sub.A at the PDM input controlled by the variable gain amplifier RV which equals its maximum value U.sub.Amax ; L.sub.p =inductance of the primary winding) EQU I.sub.pmax =(t.sub.pmax .multidot.U.sub.p)/L.sub.p, (B 3) EQU T.sub.max =t.sub.pmax /.nu. (4)
.nu. being the duty cycle defined by the equation EQU .nu.=1/(1+U.sub.p /U.multidot.U.sub.s) (5)
(u=transmission ratio of the transformer Tr)
to determine the maximum value for the secondary current I.sub.s, one obtains the equation EQU I.sub.smax =(.mu..multidot.t.sub.pmax .multidot.U.sub.p.sup.2 .multidot..nu.)/(2U.sub.s 19 L.sub.p). (6)
In the circuit of a blocking oscillator illustrated in FIG. 1, the maximum primary pulse width t.sub.pmax is determined by the current-voltage converter SSW.
In view of the circuit to be used, it is also desired that the equation EQU I.sub.p .multidot.L.sub.p =U.sub.Ip .multidot..tau..sub.ssw
be fulfilled (.tau..sub.ssw =time constant of the current-voltage converter SSR in the control circuit). This means that through its time constant .tau..sub.ssw the current-voltage converter SSW delivers a voltage U.sub.Ip proportional to the primary current I.sub.p if L.sub.p and .tau..sub.ssw remain constant.
By means of equation (3) one obtains in general: EQU T.sub.p =(I.sub.p .multidot.L.sub.p)/U.sub.p =(U.sub.Ip .multidot..tau..sub.ssw)/U.sub.p
and, for the case of the voltage U.sub.Ip delivered by the current-voltage converter SSW becoming maximum, because of EQU U.sub.Ipmax =U.sub.Amax, EQU T.sub.pmax =(U.sub.Amax .multidot..tau..sub.ssw)/U.sub.p. (7)
With the aid of equation (5) to equation (7) one obtains EQU I.sub.smax =(U.sub.Amax .multidot..tau..sub.ssw .multidot..mu.)/(2U.sub.s .multidot.L.sub.p .multidot.(1/U.sub.p)+(1/uU.sub.2))) (8)
It can be seen from equation (8) that the maximum secondary current I.sub.smax increases with increasing primary voltage U.sub.p, and that, therefore, a direct proportionality exists.
This is the above-mentioned disadvantage, which makes it necessary for high primary voltages U.sub.p to allow for secondary power ratings in the layout of the transformer Tr which in reality are not needed.
The present invention now deals with the problem of achieving a more favorable solution of the problem referred to, so as not to require an overdimensioning of the transformer Tr.